System and method for antenna diversity using equal power joint maximal ratio combining

ABSTRACT

An equal gain composite beamforming technique which includes the constraint that the power of the signal output by each antenna is the same, and is equal to the total power of the transmit signal divided by the number N of transmit antennas from which the signal is to be transmitted. By reducing output power requirements for each power amplifier, the silicon area of the power amplifiers are reduced by as much as N times (where N is equal to the number of transmit antennas) relative to a non-equal gain composite beamforming technique.A method and apparatus are disclosed for a multiple input multiple output (MIMO) transmission technique by a wireless communications device which includes providing that the power applied to each transmit antenna may be equal to the total power of the transmit signal divided by the number N of transmit antennas from which the signal is to be transmitted. The device may produce a weight for each of the N transmit antennas used in MIMO transmission. Also, the device may determine a total transmit power and produce a multi-carrier signal for transmission. The device may weight the multi-carrier signal for each antenna per the produced weight. Further, the device may apply a power to each of the N transmit antennas, for the weighted multicarrier signal, which is equal to the total transmit power divided by N. Each transmit antenna signal may be amplified by an amplifier coupled to that antenna.

Notice: More than one reissue application has been filed for the reissueof U.S. Pat. No. 7,881,674. The reissue applications are reissueapplication Ser. No. 13/755,945, which reissued as U.S. Pat. No. RE45,425 on Mar. 17, 2015, and the present application.

This application is a continuation of U.S. application Ser. No.10/800,610, filed Mar. 15, 2004, which is a continuation of U.S.application Ser. No. 10/174,689, filed Jun. 19, 2002, pending, which inturn claims priority to U.S. Provisional Application No. 60/361,055,filed Mar. 1, 2002, to U.S. Provisional Application No. 60/365,797 filedMar. 21, 2002, and to U.S. Provisional Application No. 60/380,139, filedMay 6, 2002. The entirety of each of the aforementioned applications areincorporated herein by reference.This application is a continuationreissue of U.S. patent application Ser. No. 13/755,945 filed Jan. 31,2013, which is a reissue application of U.S. patent application Ser. No.11/879,156 filed Jul. 16, 2007, which issued as U.S. Pat. No. 7,881,674on Feb. 1, 2011, which is a continuation of U.S. patent application Ser.No. 10/800,610 filed Mar. 15, 2004, which issued as U.S. Pat. No.7,245,881 on Jul. 17, 2007, which is a continuation of U.S. patentapplication Ser. No. 10/174,689 filed Jun. 19, 2002, which issued asU.S. Pat. No. 6,785,520 on Aug. 31, 2004, which claims the benefit ofU.S. Provisional Application Ser. No. 60/361,055 filed Mar. 1, 2002,U.S. Provisional Application Ser. No. 60/365,797 filed Mar. 21, 2002,and U.S. Provisional Application Ser. No. 60/380,139 filed May 6, 2002,the contents of which are hereby incorporated by reference herein.

BACKGROUND OF THE INVENTION

The present invention is directed to an antenna (spatial) processinguseful in wireless communication applications, such as short-rangewireless applications.

Composite Beamforming (CBF) is an antenna processing technique in whicha first communication device, having a plurality of antennas, weights asignal to be transmitted by its antennas to a second communicationdevice also having a plurality of antennas. Similarly, the secondcommunication device weights and combines the received signals receivedby its antennas. A multiple-input/multiple-output (MIMO) optimizedcommunication system is defined by CBF. The transmit weights and receiveweights are determined to optimize the link margin between the devices,thereby significantly extending the range of communication between thetwo communication devices. Techniques related to composite beamformingare the subject matter of above-identified commonly assigned co-pendingapplication.

There is room for still further enhancing this CBF technique to optimizecost and implementation issues at the expense of only slight degradationin performance. Such a solution is extremely valuable in manufacturing acost-effective integrated circuit solution.

SUMMARY OF THE INVENTION

An equal gain composite beamforming technique is provided that adds theconstraint that the power of the signal output by each of the pluralityof transmit antennas is the same, and is equal to the total power of thetransmit signal divided by the number N of transmit antennas from whichthe signal is to be transmitted. This reduces output power requirementsat each antenna. By reducing output power requirements for each poweramplifier, the silicon area of the power amplifiers are reduced by asmuch as N times (where N is the number of transmit antennas) relative tonon-equal gain CBF. Many implementation advantages are achieved by equalgain CBF, including savings in silicon, power requirements, etc.

The above and other objects and advantages will become more readilyapparent when reference is made to the following description taken inconjunction with the accompanying drawings.

BRIEF DESCRIPTION OF THE DRAWINGS

FIG. 1 is a block diagram of two communication devices performing equalgain composite beamforming.

FIG. 2 shows frequency dependent weights for two antennas that arefrequency shaped, but not equal gain.

FIG. 3 shows equal gain frequency dependent weights for two antennas.

FIG. 4 is a block diagram of a communication device capable ofperforming equal gain composite beamforming.

FIG. 5 is a flow diagram showing an adaptive algorithm to obtain equalgain transmit antenna weights for first and second communication devicesin communication with each other.

FIG.6FIG. 6 is a graphical diagram illustrating convergence of theadaptive algorithm shown in FIG. 5.

FIG. 7 is a graphical diagram illustrating a performance comparisonbetween equal gain composite beamforming and non-equal gain compositebeamforming.

FIG. 8 is a block diagram of a composite beamforming transmissionprocess for a multi-carrier baseband modulation scheme.

FIG. 9 is a block diagram of a composite beamforming reception processfor a multi-carrier baseband modulation scheme.

FIG. 10 is a block diagram of a composite beamforming transmissionprocess for a single carrier baseband modulation scheme.

FIG. 11 is a block diagram of a composite beamforming reception processfor a single carrier baseband modulation scheme.

FIG. 12 is a flow diagram for a process that is useful when one deviceon the communication link is composite beamforming capable and the otherdevice uses antenna selection diversity capable.

DETAILED DESCRIPTION OF THE DRAWINGS

Referring first to FIG. 1, a system 10 is shown in which a firstcommunication device and a second communication device 200 communicatewith each other using radio frequency (RF) communication techniques. Thedevices use composite beamforming techniques when communicating witheach other. In particular, communication device 100 has N plurality ofantennas 110 and communication device 200 has M plurality of antennas210. According to the composite beamforming (CBF) technique alsodescribed in the aforementioned co-pending application filed on evendate, when communication device 100 transmits a signal to communicationdevice 200, it applies to (i.e., multiplies or scales) a baseband signals to be transmitted a transmit weight vector associated with aparticular destination device, e.g., communication device 200, denotedw_(tx,1). Similarly, when communication device 200 transmits a basebandsignal s to communication device 100, it multiplies the baseband signals by a transmit weight vector w_(tx,2), associated with destinationcommunication device 100. The (M×N) frequency dependent channel matrixfrom the N plurality of antennas of the first communication device 100to M plurality of antennas of the second communication device 200 isH(k), and the frequency dependent communication channel (N×M) matrixbetween the M plurality of antennas of the second communication deviceand the N plurality of antennas of the first communication device isII^(T)(k).

The transmit weight vectors w_(tx,1) and w_(tx2) w_(tx,2) each comprisesa plurality of transmit weights corresponding to each of the N and Mantennas, respectively. Each transmit weight is a complex quantity.Moreover, each transmit weight vector is frequency dependent; it mayvary across the bandwidth of the baseband signal s to be transmitted.For example, if the baseband signal s is a multi-carrier signal of Ksub-carriers, each transmit weight for a corresponding antenna variesacross the K sub-carriers. Similarly, if the baseband signal s is asingle-carrier signal (that can be divided or synthesized into Kfrequency sub-bands), each transmit weight for a corresponding antennavaries across the bandwidth of the baseband signal. Therefore, thetransmit weight vector is dependent on frequency, or varies withfrequency sub-band/sub-carrier k, such that wtx becomes w_(tx)(f), ormore commonly referred to as w_(tx)(k), where k is the frequencysub-band/sub-carrier index.

While the terms frequency sub-band/sub-carrier are used herein inconnection with beamforming in a frequency dependent channel, it shouldbe understood that the term “sub-band” is meant to include a narrowbandwidth of spectrum forming a part of a baseband signal. The sub-bandmay be a single discrete frequency (within a suitable frequencyresolution that a device can process) or a narrow bandwidth of severalfrequencies.

The receiving communication device also weights the signals received atits antennas with a receive antenna weight vector w_(rx)(k).Communication device 100 uses a receive antenna weight vectorw_(rx,1)(k) when receiving a transmission from communication device 200,and communication device 200 uses a receive antenna weight vectorw_(rx,2)(k) when receiving a transmission from communication device 100.The receive antenna weights of each vector are matched to the receivedsignals by the receiving communication device. The receive weight vectormay also be frequency dependent.

Generally, transmit weight vector w_(tx,1) comprises a plurality oftransmit antenna weights w_(tx,1,i)=β_(1,i)(k)e^(iφ1,i,(k)), whereβ_(1,i)(k) is the magnitude of the antenna weight, φ1,i,(k) is the phaseof the antenna weight, i is the antenna index, and k is the frequencysub-band or sub-carrier index (up to K frequencysub-bands/sub-carriers). The subscripts tx,1 denote that it is a vectorthat communication device 100 uses to transmit to communication device2. Similarly, the subscripts tx,2 denote that it is a vector thatcommunication device 200 uses to transmit to communication device 100.

Under the constraint of an equal gain composite beamforming (EGCBF)process, the power of the transmit signal output by each transmitantenna is the same, and is equal to the total power associated with thetransmit signal (P_(tx)) divided by the number of transmit antennas.Thus, for communication device 100, that is P_(tx)/N. For communicationdevice 200, that is P_(tx)/M. Consequently, each power amplifierassociated with an antenna need only support 1/N of the total outputpower. Example: For N=4, P_(tx)=17 dBm, each power amplifier need onlysupport a max linear output power of 17-10log(4)=11 dBm. Thus, whereasfor non-equal gain composite beamforming each power amplifier mustsupport to the total transmit power, such is not the case for equal gainbeamforming. The equal-gain constraint makes the power amplifier designmuch simpler. Equal gain CBF performs very close to non-equal gain CBF(within 1-2 dB), but costs significantly less to implement in terms ofpower amplifier requirements and affords the opportunity to deploy thepower amplifiers on the same silicon integrated circuit as the RFcircuitry.

When considering a frequency dependent communication channel, the EGCBFconstraint implies that for each and every antenna i, the sum of thepower |w_(tx,i)(k)|² of the antenna signal across all of frequencies ofthe baseband signal (the frequency sub-bands or sub-carrier frequenciesk=1 to K) is equal to P_(tx)/N. This is the equal gain constraintapplied to a frequency dependent channel represented by K sub-carrierfrequencies or frequency sub-bands.

An additional constraint can be imposed on the frequency dependent equalgain constraint explained above. This additional constraint is afrequency shaping constraint which requires that at each frequency ofthe baseband signal to be transmitted (e.g., frequency sub-band orfrequency sub-carrier k), the sum of the power of signals across all ofthe transmit antennas (|w_(tx,i)(k)|² for i=1 to N) is equal toP_(tx)/K. This constraint is useful to ensure that, in an iterativeprocess between two communication devices, the transmit weights of thetwo devices will converge to optimal values. An additional benefit ofthis constraint is that the transmitting device can easily satisfyspectral mask requirements of a communication standard, such as IEEE802.11x.

One solution to a system that combines the frequency selective equalgain constraint and the frequency shaping constraint is that|w_(tx,i)(k)|²=P_(tx)/(KN). This solution says that the magnitude ofeach transmit antenna weight is P_(tx)/(KN). If the transmit weightvector is normalized to unity, i.e., divided by [P_(tx)/(KN)]^(1/2),then the vectors for all of the antennas across all of the k frequencysub-bands becomes a N×K matrix of phase values e^(iφik), where i is theantenna index and k is the sub-band/sub-carrier index.

$\begin{matrix}e^{j\;\varphi\; 1\; 1} & e^{j\;\varphi\; 2\; 1} & e^{j\;\varphi\; 3\; 1} & \ldots & {e^{j\;\varphi\; N\; 1}\;} \\e^{j\;\varphi\; 12} & \; & \; & \; & e^{j\;\varphi\; N\; 2} \\e^{j\;\varphi\; 13} & \; & \; & \; & e^{j\;\varphi\; N\; 3} \\\ldots & \; & \; & \; & \ldots \\e^{j\;\varphi\; 1\; K} & e^{j\;\varphi\; 2\; K} & e^{j\;\varphi\; 3K} & \ldots & e^{j\;\varphi\; N\; K}\end{matrix}$

FIG. 2 shows the magnitude of the antenna weights in a 2-antenna examplethat satisfy the frequency shaping constraint but not the equal gainconstraint. The magnitude of the antenna weights for the two antennas ateach of three exemplary frequency sub-bands (k, k+1, k+2) are shown. Thesum of the magnitude of the antenna weights of both antennas at any ofthe frequency sub-bands shown adds to a constant value, P_(tx)/K.

FIG. 3 shows the magnitude of the antenna weights in the 2-antennaexample that satisfy the equal gain constraint and the frequency shapingconstraint. In this example, the total power at antenna 1 (i.e., thetotal area under the weight curve across k frequency sub-bands) is P/2(N=2 in the example) and the total power of the antennas at anyfrequency sub-band k is P/K. Thus, the power or magnitude of thetransmit antenna weight for any given antenna is equal (constant) acrossthe bandwidth of the baseband signal, and is equal to P/(KN) for all k.In other words, at each antenna, the power is equally distributed acrossthe bandwidth of the baseband signal.

For EGCBF, the optimization problem becomesargmax{w^(H) _(tx,1)(k)H^(H)(k)H(k)w_(tx,1)(k)}, subject to|w_(tx,i)(k)|²=P_(tx)/(NK).  (1)

A closed-form solution to equation (1) is difficult to obtain since itrequires the solution of a non-linear system of equations. However, thefollowing necessary conditions for the solution to (1) have been derivedand are given below:

Optimal w_(rx) satisfies w_(rx)=kHw_(tx) for some nonzero constant k.

Optimal w_(tx) satisfies Im(Λ*H^(H)He^(iφ))=0, Λ=diag(e^(iφ0), e^(iφ1),. . . , e^(iφN-1)), w_(tx)=e^(iφ)=(e^(iφ0), e^(iφ1), . . . ,e^(iφN-1))^(T). One solution to equation (1) is an adaptive algorithmfor EGCBF. Although the algorithm is not necessarily optimal in terms ofsolving equation (1), it is simple to implement and simulations haveverified that it converges reliably at the expense of only a 1-2 dBperformance penalty relative to non-equal gain CBF. This adaptivealgorithm is described hereinafter in conjunction with FIG. 5.

The communication devices at both ends of the link , link, i.e., devices100 and 200 may have any known suitable architecture to transmit,receive and process signals. An example of a communication device blockdiagram is shown in FIG. 4. The communication device 300 comprises an RFsection 310, a baseband section 420 320 and optionally a host 330. Thereare a plurality of antennas, e.g., four antennas 302, 304, 306, 308coupled to the RF section 310 that are used for transmission andreception. The device may have only two antennas. The RF section 310 hasa transmitter (Tx) 312 that upconverts baseband signals fortransmission, and a receiver (Rx) 314 that downconverts received RFsignals for baseband processing. In the context of the compositebeamforming techniques described herein, the Tx 312 upconverts andsupplies separately weighted signals to corresponding ones of each ofthe plurality of antennas via separate power amplifiers for simultaneoustransmission. Similarly, the Rx 314 downconverts and supplies receivedsignals from each of the plurality of antennas to the baseband section320. The baseband section 320 performs processing of baseband signals torecover the information from a received signal, and to convertinformation in preparation for transmission. The baseband section 320may implement any of a variety of communication formats or standards,such as WLAN standards IEEE 802.11x, Bluetooth™, as well as otherstandards.

The intelligence to execute the computations for the compositebeamforming techniques described herein may be implemented in a varietyof ways. For example, a processor 322 in the baseband section 320 mayexecute instructions encoded on a processor readable memory 324 (RAM,ROM, EEPROM, etc.) that cause the processor 322 to perform the compositebeamforming steps described herein. Alternatively, an applicationspecific integrated circuit (ASIC) may be configured with theappropriate firmware, e.g., field programmable gates that implementdigital signal processing instructions to perform the compositebeamforming steps. This ASIC may be part of, or the entirety of, thebaseband section 320. Still another alternative is for the beamformingcomputations to be performed by a host processor 332 (in the host 330)by executing instructions stored in (or encoded on) a processor readablememory 334. The RF section 310 may be embodied by one integratedcircuit, and the baseband section 320 may be embodied by anotherintegrated circuit. The communication device on each end of thecommunication link need not have the same device architecture orimplementation.

Regardless of the specific implementation chosen, the compositebeamforming process is generally performed as follows. A transmit weightvector (comprising a plurality of complex transmit antenna weightscorresponding to the number of transmit antennas) is applied to, i.e.,scaled or multiplied by, a baseband signal to be transmitted, and eachresulting weighted signal is coupled to a transmitter where it isupconverted, amplified and coupled to a corresponding one of thetransmit antennas for simultaneous transmission. At the communicationdevice on the other end of the link, the transmit signals are detectedat each of the plurality of antennas and downconverted to a basebandsignal. Each baseband signal is multiplied by a corresponding one of thecomplex receive antenna weights and combined to form a resulting receivesignal. The architecture of the RF section necessary to accommodate thebeamforming techniques described herein may vary with a particular RFdesign, and many are known in the art and thus is not described herein.

With reference to FIG. 5, an adaptive procedure 400 for determining nearoptimum transmit antenna weight vectors for first and secondcommunication devices will be described. The antenna weight parametersin FIG. 4 are written with indexes to reflect communication between aWLAN access point (AP) and a station (STA), but without loss ofgenerality, it should be understood that this process is not limited toWLAN application, but is useful in any short-range wireless application.The AP has Nap antennas and the STA has Nsta antennas. Assuming the APbegins with a transmission to the STA, the initial AP transmit weightvector w_(T,AP,0)(k) is |1,1, . . . 1|, equal gain and normalized by1/(Nap)^(1/2) for all antennas and all frequency sub-bands/sub-carriersk. Phase for the transmit antenna weights is also initially set to zero.The index T indicates it is a transmit weight vector, index AP indicatesit is an AP vector, index 0 is the iteration of the vector, and (k)indicates that it is frequency sub-band/sub-carrier dependent. The indexi is the antenna index. The transmit weight vectors identified in FIG. 5form an N x K matrix explained above.

In step 410, a baseband signal is scaled by the initial AP transmitweight vector w_(T,A,P,0)(k), upconverted and transmitted to the STA.The transmitted signal is altered by the frequency dependent channelmatrix H(k) from AP-STA. The STA receives the signal and matches itsinitial receive weight vector w_(R,STA,0)(k) to the signals received atits antennas. In step 420, the STA gain normalizes the receive weightvector w_(R,STA,0)(k) and computes the conjugate of gain-normalizedreceive weight vector to generate the STA's initial transmit weights fortransmitting a signal back to the AP. The STA scales the signal to betransmitted to the AP by the initial transmit weight vector, upconvertsthat signal and transmits it to the AP. Computing the conjugate for thegain-normalized vector means essentially co-phasing the receive weightvector (i.e., phase conjugating the receive weight vector). Thetransmitted signal is effectively scaled by the frequency dependentchannel matrix H^(T)(k). At the AP, the receive weight vector is matchedto the signals received at its antennas. The AP then computes theconjugate of the gain-normalized receive weight vector as the nexttransmit weight vector w_(T,AP,1)(k) and in step 430 transmits a signalto the STA with that transmit weight vector. The STA receives the signaltransmitted from the AP with this next transmit weight vector andmatches to the received signals to compute a next receive weight vectorw_(R,STA,1)(k). Again, the STA computes the conjugate of thegain-normalized receive weight vector w_(R,STA,1)(k) as its nexttransmit weight vector w_(T,STA,1)(k) for transmitting a signal back tothe AP. This process repeats for several iterations, ultimatelyconverging to transmit weight vectors that achieve nearly the sameperformance as non-equal gain composite beamforming. This adaptiveprocess works even if one of the devices, such as a STA, has a singleantenna for transmission and reception. In addition, throughout theadaptive process, the frequency shaped constraint may be maintained sothat for each antenna, the transmit antenna weight is constant acrossthe bandwidth of the baseband signal.

FIG. 6 shows that the adaptive algorithm converges to performance lossthat is less than 1 dB with 95% probability after 3-4 iterations. FIG. 7shows simulation results that indicate that the performance degradationcompared to the non-equal gain composite beamforming case is only 1-2dB.

Each communication device stores the transmit antenna weights determinedto most effectively communicate with each of the other communicationdevices that it may communicate with. The transmit antenna weights maybe determined according to the adaptive algorithm described above. Whenstoring the transmit weights of a transmit weight vector, in order toconserve memory space in the communication device, the device may store,for each antenna, weights for a subset or a portion of the total numberof weights that span the bandwidth of the baseband signal. For example,if there are K weights for K frequency sub-bands or sub-carrierfrequencies, only a sampling of those weights are actually stored, suchas weights for every other, every third, every fourth, etc., k sub-bandor sub-carrier. Then, the stored subset of transmit weights areretrieved from storage when a device is to commence transmission of asignal, and the remaining weights are generated by interpolation fromthe stored subset of weights. Any suitable interpolation can be used,such as linear interpolation, to obtain the complete set of weightsacross the K sub-bands or sub-carriers for each antenna.

With reference to FIG. 8, a beamforming transmission process 500 isshown for a multi-carrier baseband modulation scheme. For amulti-carrier modulation system, such as an orthogonal frequencydivision multiplexed (OFDM) system used in IEEE 802.11a, the datasymbols are in the frequency domain. K symbols are assigned to Ksub-carriers (K=52 for 802.11a). For convenience, each of the transmitantenna weights are described as a function of (k), the sub-carrierfrequency. The equal gain transmit antenna weights are computed by anyof the processes described herein at each of the sub-carrierfrequencies. There is a signal processing path for each of the Nantennas. In each signal processing path, a multiplier 510 multipliesthe frequency domain symbol s(k) by the corresponding equal-gaintransmit antenna weight w_(tx)(k) and because w_(tx)(k) has K values,there are K results from the multiplication process. The results arestored in a buffer 520 for k=1 to K. An inverse Fast Fourier Transform(IFFT) 530 is coupled to the buffer to convert the frequency domainresults stored in buffer 520 to a digital time domain signal for each ofthe K sub-carriers. There may be some adjustment made for cyclicprefixes caused by the OFDM process. A filter 540 provides lowpassfiltering of the result of the IFFT process. The digital results of thefiltering process are converted to analog signals by a D/A 550. Theoutputs of the D/A 550 are coupled to RF circuitry 560 that upconvertsthe analog signals to the appropriate RF signal which is coupled via apower amplifier (PA) 570 to one of the N antennas 580. In this manner,for each antenna 580, the signal s(k) is multiplied by respectivetransmit antenna weights whose phase values may vary as a function ofthe sub-carrier frequency k.

FIG. 9 shows a reception process 600 that is essentially the inverse ofthe transmission process 500 shown in FIG. 8. There is a signalprocessing channel for each of the antennas 580. RF circuitry 610downconverts the RF signals detected at each antenna 580 for each of thesub-carriers. An A/D 620 converts the analog signal to a digital signal.A lowpass filter 630 filters the digital signal. There may be someadjustment made for cyclic prefixes caused by the OFDM process. A buffer640 stores the time domain digital signal in slots associated with eachsub-carrier frequency k. An FFT 650 converts the time domain digitalsignal in buffer 640 to a frequency domain signal corresponding to eachsub-carrier frequency k. The output of the FFT 650 is coupled to amultiplier 660 that multiplies the digital signal for each sub-carrier kby a corresponding receive antenna weight w_(rx)(k) for thecorresponding one of the N antennas. The outputs of each of themultipliers 660 are combined by an adder 670 to recover the digitalfrequency domain symbol s(k). The signal s(k) is then mapped back tosymbol b(k).

FIGS. 10 and 11 show transmission and reception processes, respectively,that are applicable to a single-carrier baseband modulation scheme, suchas that used by the IEEE 802.11b standard. The data symbols in such asystem are m in the time domain. FIG. 10 shows a beamformingtransmission process 700. Since in a frequency dependent channel, thetransmit antenna weights are frequency dependent, the passband of thetime-domain baseband signal is synthesized into frequency bins orsub-bands (K bins or sub-bands) and equal gain transmit beamformingweights are computed for each frequency bin using any of the processesdescribed herein. There are processing channels for each antenna. Ineach processing channel, transmit filters 710 are synthesized with thefrequency response specified by the beamforming weights. Thus, eachtransmit filter 710 has a frequency response defined by the transmitantenna weight w_(tx)(f) associated with that antenna. The data symbols(n) is passed through the transmit filter 710 which in effect appliesthe frequency dependent antenna weight w_(tx)(f) to the data symbols(n). The D/A 720 converts the digital output of the transmit filter 710to an analog signal. The RF circuitry 730 upconverts the analog signaland couples the upconverted analog signal to an antenna 750 via a poweramplifier 740.

FIG. 11 shows a reception process 800 suitable for a single carriersystem. There is a processing channel for each antenna 750. In eachprocessing channel, RF circuitry 810 downconverts the received RFsignal. An A/D 820 converts the downconverted analog signal to a digitalsignal. Like the frequency dependent transmit antenna weights, thereceive antenna weights are computed for the plurality of frequencysub-bands. Receive filters 830 are synthesized with the frequencyresponse specified by the frequency dependent receive beamformingweights w_(rx)(f) and the received digital signal is passed throughfilters 830 for each antenna, effectively applying a frequency dependentantenna weight to the received signal for each antenna. The results ofthe filters 830 are combined in an adder 850, and then passed to ademodulator 860 for demodulation.

FIG. 12 shows a procedure for use when only one of the two devicessupports beamforming. For example, N-CBF is supported at a firstcommunication device (an AP) but not at a second communication device (aSTA). In this case, the STA is likely to support 2-antenna Tx/Rxselection diversity as discussed previously. If this is the case, it ispossible for the AP to achieve 3 dB better performance than Nth ordermaximal ratio combining (MRC) at both ends of the link.

When a STA associates or whenever a significant change in channelresponse is detected, the AP sends a special training sequence to helpthe STA select the best of its two antennas. The training sequence usesmessages entirely supported by the applicable media access controlprotocol, which in the following example is IEEE 802.11x.

The sequence consists of 2 MSDUs (ideally containing data that isactually meant for the STA so as not to incur a loss in throughput). Instep 900, the AP sends the first MSDU using a Tx weight vector havingequal gain orthogonal transmit weights (also optionally frequencyshaped). For example, if the number of AP antennas is 4, the transmitweight vector is [1 1 1 1]^(T). In step 910, the 2-antenna selectiondiversity STA responds by transmitting a message using one of its' itstwo antennas; the antenna that best received the signal from the AP. TheAP decodes the message from the STA, and obtains one row of the H matrix(such as the first row h_(r1)). In step 920, the AP sends the secondMSDU using a frequency dependent transmit weight vector which isorthogonal to the first row of H (determined in step 610 910), andequal-gain normalized such that the magnitude of the signals at eachantenna is equal to P/N. In addition, the transmit weight vector may befrequency shaped across so that at each frequency of the basebandsignal, the sum of the power output by the antennas of the firstcommunication device is constant across. When the STA receives thesecond MSDU, it forces the STA to transmit a response message in step630 930 using the other antenna, allowing the AP to see the second rowof the H matrix, such as h_(r2). Now the AP knows the entire H matrix.The AP then decides which row of the H matrix will provide “better” MRCat the STA by computing a norm of each row, h_(r1) and h_(r2), of the Hmatrix and, and selecting the row that has the greater norm as thefrequency dependent transmit weight vector for further transmissions tothat STA until another change is detected in the channel. The row thatis selected is equal gain normalized before it is used for transmittingto that STA.

Equal gain composite beamforming provides significant advantages. Byreducing output power requirements for each power amplifier, the siliconarea of the power amplifiers are reduced by as much as N times (where Nis equal to the number of antennas) relative to non-equal gain CBF. Thesilicon area savings translates into a cost savings due to (1) lesssilicon area, and (2) the ability to integrate the power amplifiers ontothe same die (perhaps even the same die as the radio frequencytransceiver itself).

The efficiency (efficiency being defined as the output power divided byDC current consumption) for each power amplifier is N times larger inthe EGCBF case than in the non-equal gain CBF case. With EGCBF, the sameoutput power as non-equal CBF is achieved with N times less DC current.

Equal gain CBF reduces power requirements for each of the poweramplifiers, which in turn increases the output impedance of each of thepower amplifiers (since impedance is inversely proportional to current,and supply current requirements will be reduced). When the outputimpedance of the power amplifier is increased, the matching networksrequired for the power amplifiers are greatly simplified and requireless silicon area. Moreover, reducing power requirements for theindividual power amplifiers provides greater flexibility for systemswith low supply voltage.

To summarize, a method is provided that accomplishes applying a transmitweight vector to a baseband signal to be transmitted from the firstcommunication device to the second communication device, the transmitweight vector comprising a complex transmit antenna weight for each of Nplurality of antennas of the first communication device, wherein eachcomplex transmit antenna weight has a magnitude and a phase whose valuesmay vary with frequency across a bandwidth of the baseband signal,thereby generating N transmit signals each of which is weighted acrossthe bandwidth of the baseband signal, wherein the magnitude of thecomplex transmit antenna weight associated with each antenna is suchthat the power to be output at each antenna is the same and is equal tothe total power to be output by all of the N antennas divided by N;receiving at the N plurality of antennas of the first communicationdevice a signal that was transmitted by the second communication device;determining a receive weight vector comprising a plurality of complexreceive antenna weights for the N plurality of antennas of the firstcommunication device from the signals received by the N plurality ofantennas, wherein each receive antenna weight has a magnitude and aphase whose values may vary with frequency; and updating the transmitweight vector for the N plurality of antennas of the first communicationdevice for transmitting signals to the second communication device bygain normalizing the receive weight vector and computing a conjugatethereof. This process may be performed such that at substantially allfrequencies of the baseband signal, the sum of the magnitude of thecomplex transmit antenna weights across the plurality of antennas of thefirst communication device is constant. Moreover, where the bandwidth ofthe baseband signal comprises K plurality of frequency sub-bands, themagnitude of the complex transmit antenna weights associated with eachof the N plurality of antennas is such that the power to be output byeach antenna is the same and is equal to 1/(KN) of the total power to beoutput for all of the K frequency sub-bands. This process may beembodied by instructions encoded on a medium or in a communicationdevice.

In addition, a method is provided that accomplishes a method forcommunicating signals from a first communication device to a secondcommunication device using radio frequency (RF) communicationtechniques, comprising: applying to a first signal to be transmittedfrom the first communication device to the second communication device atransmit weight vector, the transmit weight vector comprising a complextransmit antenna weight for each of the N plurality of antennas, whereineach complex transmit antenna weight has a magnitude and a phase whosevalues may vary with frequency across a bandwidth of the basebandsignal, thereby generating N transmit signals each of which is weightedacross the bandwidth of the baseband signal, wherein the magnitude ofthe complex transmit antenna weights associated with each antenna issuch that the power to be output at each antenna is the same and isequal to the total power to be output by all of the N antennas dividedby N; from a first response signal at the plurality of antennas of thefirst communication device transmitted from a first of two antennas ofthe second communication device, deriving a first row of a channelresponse matrix that describes the channel response between the firstcommunication device and the second communication device; applying to asecond signal for transmission by the plurality of antennas of the firstcommunication device a transmit weight vector that is orthogonal to thefirst row of the channel response matrix, and wherein the transmitweight vector comprises a plurality of complex transmit antenna weights,wherein each complex transmit antenna weight has a magnitude and a phasewhose values may vary with frequency across a bandwidth of the secondsignal, thereby generating N transmit signals each of which is weightedacross the bandwidth of the baseband signal, wherein the magnitude ofthe complex transmit antenna weights associated with each antenna issuch that the power to be output at each antenna is the same and isequal to the total power to be output by all of the N antennas dividedby N; deriving a second row of the channel response matrix from a secondresponse signal received from a second of the two antennas of the secondcommunication device; and selecting one of the first and second rows ofthe channel response matrix that provides better signal-to-noise at thesecond communication device as the transmit weight vector for furthertransmission of signals to the second communication device. This processmay be performed such that at substantially all frequencies of thebaseband signal, the sum of the magnitude of the complex transmitantenna weights across the plurality of antennas of the firstcommunication device is constant. Moreover, where the bandwidth of thebaseband signal comprises K plurality of frequency sub-bands, themagnitude of the complex transmit antenna weights associated with eachof the N plurality of antennas is such that the power to be output byeach antenna is the same and is equal to 1/(KN) of the total power to beoutput for all of the K frequency sub-bands. This process may beembodied by instructions encoded on a medium or in a communicationdevice.

The above description is intended by way of example only.

What is claimed is:
 1. A wireless communication device, comprising: aplurality of N antennas; a baseband processor configured to determine areceive weight vector of a plurality of complex receive antenna weightsfor each of the plurality of N antennas, the receive antenna weightsapplied to a received baseband signal; compute a transmit weight vectorby computing a conjugate of the receive weight vector, the transmitweight vector comprising a complex transmit antenna weight for each ofplurality of N antennas of the communication device, wherein eachcomplex transmit antenna weight has a magnitude and a phase whose valuesmay vary with frequency across a bandwidth of the baseband signal,thereby generating a plurality of N transmit signals each of which isweighted across the bandwidth of the baseband signal to be transmittedfrom corresponding ones of the plurality of N antennas to a secondcommunication device, wherein the magnitude of the complex transmitantenna weight associated with each antenna is such that the power to beoutput at each antenna is the same and is equal to the total power to beoutput by all of the N antennas divided by N and such that the sum ofthe power at each corresponding frequency across the plurality oftransmit signals is equal to a constant; apply the transmit weightvector to a baseband signal for transmission via the plurality of Nantennas; and update the transmit weight vector by repeating thedetermining of the receive weight vector and computing of the transmitweight vector each time signals are received to update the transmitweight vector.
 2. The device of claim 1, wherein the bandwidth of thebaseband signal comprises K plurality of frequency sub-bands, and themagnitude of the complex transmit antenna weights associated with eachof the plurality of N antennas is such that the power to be output byeach antenna is the same and is equal to 1/(KN) of the total power to beoutput for all of the K frequency sub-bands of the communication device.3. The device of claim 2, further comprising a baseband memoryconfigured to store, for each of the N antennas, complex transmitantenna weights for a subset of the K frequency sub-bands orsub-carriers.
 4. The device of claim 3, wherein the baseband processorand the stored subset of complex transmit antenna N and generatetherefrom the complete set of antenna weights for all of the K frequencysub-bands or sub-carriers using interpolation techniques.
 5. The deviceof claim 1, wherein the receive weight vector, the transmit weightvector and the baseband signal of the are applied to each of K frequencysub-bands of the baseband signal that correspond to sub-carriers of amulti-carrier baseband signal or synthesized frequency sub-bands of asingle carrier baseband signal.
 6. A wireless communication devicecomprising: a plurality of N antennas; and a processor configured toproduce a weight for each of the plurality of N antennas for use inmultiple input multiple output (MIMO) transmission; wherein theprocessor is further configured to determine a total transmit power;wherein the processor is further configured to produce at least onemulti-carrier signal; and wherein the processor is further configured toweight the at least one multi-carrier signal for each of the N antennasper the produced weight; a transmitter, operatively coupled to theprocessor, the transmitter and the processor configured to apply a powerto each of the N antennas, for the at least one weighted multi-carriersignal, equal to the total transmit power divided by N; and thetransmitter, operatively coupled to the plurality of N antennas, thetransmitter and the plurality of N antennas configured to transmit theat least one weighted multi-carrier signal.
 7. The wirelesscommunication device of claim 6 wherein an amplifier is coupled to eachof the antennas.
 8. The wireless communication device of claim 6 whereinthe multi-carrier signal is an orthogonal frequency division multiplexsignal.
 9. The wireless communication device of claim 6 wherein themulti-carrier signal has a plurality of K subcarriers; and wherein apower applied to each of the K subcarriers per antenna is equal to thetotal transmit power divided by KN.
 10. A method comprising: producing,by a wireless communication device, a weight for each of a plurality ofN antennas for use in multiple input multiple output (MIMO)transmission; determining, by the wireless communication device, a totaltransmit power; producing, by the wireless communication device, atleast one multi-carrier signal; weighting, by the wireless communicationdevice, the at least one multi-carrier signal for each of the N antennasper the produced weight; and applying, by the wireless communicationdevice, a power to each of the N antennas, for the at least one weightedmulti-carrier signal, equal to the total transmit power divided by N;and, transmitting, by the wireless communication device, the at leastone multi-carrier signal.
 11. The method of claim 10 further comprising:for each antenna, amplifying a signal, by an amplifier coupled to thatantenna.
 12. The method of claim 10 wherein the multi-carrier signal isan orthogonal frequency division multiplex signal.
 13. The method ofclaim 10 wherein the multi-carrier signal has a plurality of Ksubcarriers; and wherein a power applied to each of the K subcarriersper antenna is equal to the total transmit power divided by KN.